Multichannel absorberless near field measurement system

ABSTRACT

A near field microwave scanning system includes a switched array of antenna elements forming an array surface, a scan surface substantially parallel to the array surface and separated by a distance less than about 1 wavelength of the measured frequency, and a processing engine for obtaining and processing near field data, without the use of an absorber.

FIELD OF THE INVENTION

The present invention is directed to the measurement, testing andverification of performance parameters of electromagnetic radiatingdevices.

BACKGROUND OF THE INVENTION

Performance parameters of electromagnetic radiating devices includeEffective Isotropic Radiated Power (EIRP) and Effective Radiated Power(ERP), radiation pattern, directivity, RF current distribution ofmounting surfaces and magnetic near field distribution. Such radiatingdevices may include multimode, multiband, or multiple input/multipleoutput (MIMO) radiating devices, such as for example, cellular phones,and wireless transceivers including WiFi gear, and wireless PDA's andlaptop computers.

When cellular phones or other radiating devices are manufactured, theymust be calibrated to transmit a known RF power (galvanic power) fromthe transmitter to the antenna structure, as well as to radiate known RFpower (EIRP/ERP) from the antenna structure. The power measurement andverification also must be performed at various levels throughout theoperating range of the radiating device. This measurement andverification ensures that the highest power transmitted to and from theantenna produces legal and acceptable specific absorption rate (SAR),for a given EIRP/ERP limit. As well, the power measurement andverification assists in maintaining a valid wireless link in cellularcommunications while minimizing power usage, thereby extending batterylife, and maximizing coverage and capacity of the cell sector.

Conventionally, a sample of every cell phone model to be retail marketedis tested for the maximum EIRP/ERP Level in a test lab for severalhours, with a considerably large measurement uncertainty of greater than2.0 dB. Before performing this test, the galvanic power of the cellphone must be calibrated and the cell phone is set to radiate withmaximum galvanic power.

The cell phone RF power is conventionally delivered to the cell phonetest set using a physical hardwired connector just before the antennasection of the RF circuit, and is adjusted via a cable connectionbetween the RF connector on the cell phone and test set. Once themaximum EIRP/ERP level is adjusted or found to meet regulatory limitsfor a given galvanic power, then only the SAR level measurements areperformed for legal compliance.

To measure and verify the RF power of a cell phone or other radiatingdevices having more than one antenna, as well as for devices with MIMOarchitecture, the manufacturer usually provides a single RF connectoralong with an RF switch, filter and impedance matching for each antennacircuit. As the RF connector is well before the RF switch, filter andmatching circuits, the performance of each of the antenna circuit isunknown even after successfully completing all the manufacturing testsof the cell phone using the conventional method.

While performing the SAR measurement, the maximum galvanic power levelobtained in the first step is used as the starting level. If thegalvanic power requires adjustment to meet the SAR limits, the adjustedgalvanic power level will be considered the maximum power that can befed to the antenna, and then EIRP/ERP levels must be re-evaluated.

Most manufactured cell phone (or radiating device) samples of the samemodel are calibrated using the new galvanic power level as the maximumpower to the antenna. Once this maximum level is measured and verified,up to 20 intermediate power levels are set and measured throughout thedynamic range. In order to perform these measurements, a galvanic RFlink is established using a cable between the cell phone RF connectorand the test set. The RF connector of the cable for the cell phoneconnection end wears out over time and is replaced based on an estimatedmaximum number of insertions during the manufacturing test cycle for allproduced units (usually very large). Production testing is stopped and anew cable must be introduced and a recalibrated before manufacturingtesting can resume. This introduces delay and cost.

After each cell phone is measured and verified for appropriate galvanicRF power levels in order to meet the legal EIRP/ERP as well as SARlevels, each cell phone is further tested for Tx and Rx performance. Toperform this test, the cell phone is connected to the cell phone testerusing an RF cable between its RF connector and test equipment asdiscussed above. In the majority of the cases, the RF power measurementand verification is done in one location and the Tx/Rx parametrictesting is done in another location. In the event these tests areperformed at different locations, the RF cable connected between thecell phone and test set must be replaced frequently with a new RF cabledue to the large number of insertions. A recalibration of the RF cablemust be performed before continuing the manufacturing Tx/Rx parametrictesting of cell phones which introduces further delay and cost.

During board level manufacturing or designer testing to optimize the RFparameters of cell phones, the measurements are performed with an RFgalvanic connection. This method does not provide all the necessarymeasurements to understand the complete performance of the RF circuit.

During the design and development of radiating devices, designers oftengo though a series of iterations to improve the radiated performance ofthe antenna model(s) for achieving greater usable range, both infrequency and sensitivity, while targeting low SAR levels and lowgalvanic RF power. Each time the radiated performance of the radiatingdevice is measured, it is necessary to go to the test labs whereEIRP/ERP levels can be optimized through a series of measurements.Currently no tool exists for finding accurate spatial distribution ofthe RF radiation in the near field to minimize unwanted radiation.Designers rely on the conventional testing methods in the test labs forfar field radiated patterns and then debug at the circuit board level,which is a very tedious and complicated process.

For measuring antenna properties such as radiation pattern, gain, anddirectivity, near field scanners are employed to gather accurateamplitude and phase data and subsequently to calculate the equivalentfar field value using one of many transformations known and available inthe prior art. To accurately estimate the far field, those skilled inthe art believe the measurement distance between the probe and antennaunder test should be greater than or equal to one wavelength. Currentnear field testing is performed using a mechanical scanner with a singlecompensated probe which can detect both polarizations. Thesemeasurements usually take more than a few hours to complete a scan ofthe entire radiating surface.

When near field radiation is measured, the array elements and conductingplanes and dielectric medium around them have considerable effect on thenear field distribution of the radiating source as well as its far fieldproperties. In the prior art, using multi-axis near field measurementsystems, the measurement is performed at greater than one wave lengthfrom the antenna under test in order to minimize the ground plane effectwhich is then accounted for relatively easily. The array sensitivity isdecreased and the measurement dynamic range is limited. Furthermore,measurement speed and physical size make this an impractical approach ina high speed production test environment or in a traditional developmentlab where real time feedback and the effective use of physical lab spaceis highly valued.

In another approach, a perfect near field absorber such as thatdescribed in U.S. Pat. No. 6,762,726 B2, issued Jul. 13, 2004, is usedto increase the isolation between the radiating and array surfaces thusdecreasing the mutual coupling effects which will distort the measuredfield strength of the electromagnetic radiation emitted from thecircuitry transmitting the signal. The array sensitivity issignificantly decreased, and the measurement dynamic range is limited.Furthermore, the probe density described, and the required attributesand performance of the added physical absorber solution add tremendouscomplexity, sustainable yield challenges, and cost to deploying aphysically realizable solution. With the addition of a physicalabsorber, the interaction between radiating source and the absorbersurface still exists and results in a modified near-field representationof the radiating source.

There is a need in the art for method and apparatus of measuringperformance such as EIRP and ERP from electromagnetic radiating devicesusing near field measurement techniques that addresses the limitation ofthe solutions referenced.

SUMMARY OF THE INVENTION

The present invention comprises a novel multi-channel near fieldscanning system for measuring performance parameters such as EIRP/ERPand generating far field patterns of electromagnetic radiating devicesthrough a range of input power levels. Preferably, the scanning systemis also transparent for accurate and repeatable measurement ofroundtripTx/Rx performance. In at least one embodiment, this system iseffective without the need for a galvanic RF connection. A radiatingdevice such as a multimode, or multiband, or MIMO (or combinationsthereof) mobile or cellular phone is placed on a scanner of finite areaat distances equal to or smaller than about 1/1.8^(th) of a wavelengthof the operating frequency of the radiating source. Preferably, thedistance is about 1/1.8^(th) to about 1/88^(th) of the wavelength, overa frequency range of about 8 GHz to about 170 MHz. A multi-channelelectromagnetic scan is performed in real time using an electronicallyswitched array of probes, and the near field amplitude and phase of boththe x and y components of the radiating source are measured, corrected,re-measured and displayed. Using the corrected near field data, farfield transformations and radiating source models, performanceparameters such as the EIRP/ERP, directivity and radiation pattern ofthe radiating device are estimated and displayed.

Due to its real time scanning speeds and accurate near and far fieldmeasurement capability, embodiments of the present invention may be usedto rapidly test in a production environment or to characterize theradiating devices, to measure the RF current distribution on mountingsurfaces of antennas, to improve RF circuits, to debug and locate faultyantennas or sub-arrays or arrays and to optimize antenna performance.

Embodiments of the present invention may also measure the Tx./Rxperformance of a radiating source without the need for a galvanic RFconnection. The radiating device, which may be a multimode and/ormultiband and/or MIMO mobile or cellular phone, is placed on the scannerat distances smaller than one wavelength, preferably 1/1.8^(th)-1/88^(th) of one wavelength, of the operating frequency of the radiatingsource. Two distinct and optimum RF channels of the near field scannerare selected and are assigned to Tx and Rx modes of the transceiver.Using an external test set, the Tx and Rx performance of transceiver isevaluated.

Therefore, in one aspect, the invention may comprise an absorberlessmultichannel near field microwave scanning system comprising:

-   -   (a) a switched array of antenna elements embedded in a        dielectric, for sensing electromagnetic field components at        pre-determined locations, and forming an array surface, wherein        said array outputs raw uncorrected signals which are indicative        of the electromagnetic field and which include mutual coupling        effects;    -   (b) a scan surface for placement of the device under test (DUT),        wherein the scan surface is substantially parallel to the array        surface and separated by a distance less than about 1/1.8^(th)        of the wavelength of the measured frequency;    -   (c) a processing engine operatively connected to the switched        antenna array for obtaining and processing the array output,        said processing engine adapted to correct at an individual probe        level for mutual coupling effects.

In one embodiment, the mutual coupling effects include reflections anddynamic coupling effects between individual antenna elements across thearray, and close array to DUT proximity effect. As well, the finitescanner size may also affect the far field transformation and isaccounted for in the processing engine.

In one embodiment, the processing engine comprises:

-   -   i. a controller,    -   ii. a channel selector and sampler,    -   iii. a channel corrector, to precisely adjust for differential        path losses and delays;    -   iv. a data translator and interpolator,    -   v. an amplitude and phase detector,    -   vi. a near field corrector, to correct at an individual probe        level for reflections, and dynamic coupling between individual        antenna elements across the array    -   vii. a transformer for transforming the near field data to far        field patterns and design performance parameters, and    -   viii. a user interface.

In another aspect, the invention may comprise an absorberless method ofmeasuring EIRP/ERP or Tx/Rx performance of RF and microwave transceiver,said method comprising the steps of:

-   -   (a) using a switched array of antenna elements forming an array        surface;    -   (b) using a scan surface, wherein the scan surface is        substantially parallel to the array surface and separated by a        distance less than about one half wavelength of the measured        frequency;    -   (c) generating near field data by receiving output from each        antenna which is indicative of an electromagnetic field but        includes mutual coupling effects and effects due to finite        scanner size;    -   (d) correcting near field data to correct at an individual probe        level for reflections and mutual coupling effects across the        array; and    -   (e) transforming the near field data to far field data.

In one embodiment, the mutual coupling effects include reflections anddynamic coupling effects between individual antenna elements across thearray and the close array to DUT proximity effect.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will now be described by way of an exemplary embodimentwith reference to the accompanying simplified, diagrammatic,not-to-scale drawings. In the drawings:

FIG. 1 is a schematic representation of the antenna array of a scanner.

FIG. 2 is a side view of the antenna array and the scan plane.

FIG. 3 shows alternative arrangements of a half-loop antenna array.

FIG. 4 shows a two-layer switch array.

FIG. 5A shows a schematic depiction of the processing engine, and FIG.5B shows a schematic representation of the controller function. FIG. 5Cshows a schematic flowchart depicting near field correction.

FIG. 6 shows a schematic depiction of the exterior fields of a radiatingantenna.

FIG. 7 shows a schematic depiction of the geometry of a planar nearfield measurement.

FIGS. 8A-8E show different screenshots of various displays produced bythe graphical user interface of the processing engine.

DETAILED DESCRIPTION OF THE INVENTION

The present invention provides for a method and apparatus for measuringradiated power of a radiating source in the near field. When describingthe present invention, all terms not defined herein have their commonart-recognized meanings. The term “about” when used in combination witha numerical value, shall mean the value includes a range of 10% aboveand below the stated value or within the known tolerances of the methodsof measuring the value. The term “near field” means the field within adistance from the antenna less than or equal to about one wavelength ofthe radio frequency being radiated. Where permitted, the referenceslisted herein are incorporated herein as if reproduced in theirentirety.

The present invention comprises an absorber-less microwave near fieldscanner. In one embodiment, the scanner (100) includes a plurality ofantennas (101) arrayed in a two-dimensional array capable oftransmitting or receiving electromagnetic radiation. The antennas arepreferably, but not necessarily, half loop antennas. As shown in FIG. 1,the array may have m elements in the x-axis, and n elements in they-axis. In one embodiment, the loop dimensions length (L) and depth areoptimized to provide sufficient discrimination between H-Field intensityand E-Field intensities. For a given scan area and radiated poweraccuracy, the inter element spacing (d) and the total number of arrayelements are determined. In one embodiment, m may be 24 while n may be16, while d may equal about 10 mm. In one embodiment, d may equal about5 L. A greater number of antennas within given area (ie. smaller valuesof d) will provide greater accuracy, however, at a cost of increasedmutual coupling effects between antennas and their feed structures.

As shown in FIG. 2A, the scan plane (102) is placed at a distance (D)from the array surface (103) with the preferred range of about 1/88^(th)to about 1/1.8^(th) of wavelength and the corresponding inter elementdistance (d) range of the array is preferably about 1/176^(th) to about1/3.6^(th) of wavelength. If D is such that the scan plane is too closeto the array surface, the array surface may be within the very reactivenear field, as shown in the FIG. 6, with negative consequences. However,as D gets larger, the size of the array must be increased to obtain thesame scan energy. In one embodiment, D/d may be about 2.0.

The embodiment illustrated shows a substantially planar scan plane andantenna array, which are substantially parallel to each other.Alternative embodiments may include spherical, cylindrical or othergeometrical scan surfaces.

The typical layout of the half loops (101) is such that the successivearray elements transmit or receive the orthogonal polarizations of Hfield intensity. Alternative layout arrangements can also be used withthis scanner or array, including those arrangements shown in FIG. 3.

The outputs from the array antennas are fed through a backplane to thesecondary side of a multilayer printed circuit board (PCB). The PCBlayer stack and layout should preferably be able to provide anelement-to-element isolation better than 20 dB over the frequency rangeof interest. In one embodiment, the one end of the half loop antenna(101) is connected to a ground plane immediately adjacent the antennalayer, and other end of the half loop antenna is connected to amicrostripline layer through a feed via, without matching.

The output from a specific antenna (101) is selected by means ofswitches (110) which can select the output from any one of the antennaelements (101). Because the large number of antenna elements makes asingle switch for each antenna difficult to implement, one embodiment ofthe invention comprises a system of layered switches, which permits theuse of a relatively small number of switches. In one example, using 3layers of SP4T switches, the number of signals can be reduced by afactor of 64. Thus, a 384 element array can be reduced to 6 RF outputs.A module of 16 elements with two layers of switches is in FIG. 4.

Along with the switch matrix, a second channel is connected to oneantenna element to provide a reference signal. This reference signal isnecessary to make relative phase measurements. The architecture ofantenna array is such that it is expandable to simultaneous radiatedpower measurement of more than one radiating surface or device byappropriately selecting a pair of channels.

A selected and a non-selected antenna should preferably be adequatelyisolated from each other. Poor isolation phenomena is typically due toleakage in the cavity formed by adjoining ground planes, such that theantenna feed vias extend conduction of the antenna currents through theinner layers to the component layer, making the feed via an effectiveradiator.

Measurement and simulation of antenna isolation in a two layer boardshowed very good isolation. As a result, we believe that the couplingwas not due to the antenna structure and was not present with only oneground layer. When the simulation model was changed to include more thantwo ground layers, significant leakage was observed. Power would flowfrom one via to the next in the substrate between the ground layers withvery little attenuation.

A coaxial feed may provide excellent isolation in theory, however, acoaxial feed may be difficult to manufacture. A more practical solutionmay be to implement with ground via or ground ties. Therefore, in oneembodiment, the PCB includes isolation means consisting of a ground via(or a ground tie) connecting all the ground layers together. We havefound that positioning the ground tie closer to the feed via producesbetter isolation, and using multiple ground ties also produces betterisolation.

The processing engine accepts antenna signals from the scanner PCB,referred to herein as near field data, and processes them to provideuseful information. The antenna signals include mutual coupling effectssuch as reflections and dynamic coupling between individual antennaelements across the array, and the proximity of the device-under-test(DUT) to the array, as well as effects related to the finite scannersize, both physical and virtual. Therefore, in one embodiment, theprocessing engine provides means for removing or minimizing mutualcoupling effects at the individual probe level. The processing enginealso accounts for the close array to DUT proximity effect, and furtheraccounts for a finite virtual scanner size through the transformation tofar field using a plane wave spectrum (PWS) model.

In one embodiment, as shown schematically in FIG. 5, the processingengine (10) comprises a controller (12), a channel selector and sampler(14), a channel corrector (16) to precisely adjust for differential pathlosses and delays, a data translator and interpolator (18), an amplitudeand phase detector (20), a near field corrector (22), a transformer fortransforming the near field data to far field data (24), and a userinterface including a graphics card or other means for driving a display(26). The processing engine may also include a post-processor (28) andmeans for determining EIRP (30). A traceability module (32) is optional.The components of the processing engine may be implemented in software,firmware, hardware, or any combination thereof, as is well-known in theart.

As shown in FIG. 5B, the controller (12) functions primarily to supplypower to the rest of the system and to control the switches andattenuators on the PCB. The controller (12) receives commands from theoperating processor, which may be a desktop or laptop computer, andtranslates this data into the signals required to operate the antennaboard and the signal conditioning section. The control board inputs areconnected to the I/O on the computer. The input lines are used for datatransfer of state control signals as well as routing signals, which willcause the proper state control to go to the proper output data lines.

To have complete control over the state of the antenna board and thesignal conditioning system, and to obtain accurate measurements acrossthe required dynamic range, two sets of inputs are required. One of thebits on the input is dedicated to the group select of the input. Thefeedback and delay section is necessary to handle the handshakingrequirements of the I/O card. This section is also used to generate theCLK signal.

The ACK signal is sent from the I/O card and requires a REQ to bereturned before the card will output the next set of data. The REQsignal must have a certain minimum delay and duration. This handshakingrequirement is met by a simple feedback and delay circuit. The REQ delayis introduced by passing the ACK signal through two invertersimplemented using NAND gates. A CLK signal is also introduced into thesystem using the two inverter approach.

Since the total output data lines required from the power and controlboard are many (38 lines in the embodiment shown), some form ofde-multiplexing or decoding is needed and in a preferred embodiment,both strategies are employed on the board. Suitable demultiplexing anddecoding strategies are well-known to those skilled in the art.

The user interface and display (22) may display data on a conventionalcomputer monitor, and accept user inputs via a computer keyboard, andmouse, as is well known in the art. In one embodiment, the userinterface is a graphical user interface (GUI) and the displayarchitecture is designed to provide flexibility to feed test parameterssuch as selection of the scan area, reference probes, scan type,selection of models, frequency range and loading the data such as rawdata, DAQ corrected data, probe corrected data, translated data, pathcorrected data and reference far field data. Once all test parametersare loaded, the GUI and display section (22) of the processing engine(10) interprets the loaded test parameters and create a test sequenceand starts executing each test sequence with the help of the controllerwhile simultaneously measuring/logging the scan data to a computermemory. Additionally or optionally, the scan data may be written to ahard drive or other data storage device for further processing.

The scan data is then further processed to determine at least oneperformance parameter such as the 2D and 3D component specific nearfield distribution, total near field distribution, amplitude and phasedistribution, far field pattern in principle cuts and for any desiredcuts, as well as ERP, EIRP and directivity in real time.

In a two channel system, the channels are designated as reference andmeasurement channels respectively. In one embodiment, the referencechannel is connected to an unique element of the array, however, it canalso be reconfigured to connect to different elements of the array whichmay be determined dynamically by the controller, based on the scannedinformation or input parameters. In one embodiment, the systemarchitecture enables to select a pair of antenna elements of the arrayand connects them to reference and measurement channels simultaneously.

In one embodiment, both the reference and measurement input channels aremixed down to an intermediate frequency (IF). The IF signals are furtheramplified and processed through band pass filters. These filters willdetermine the frequency range of the IF, so in order to cover the fullmeasurement frequency range, the local oscillator (LO) will need to beprogrammed to generate the correct IF range. The full input frequencyrange will be broken down into N segments equal in width to the IFfilter bandwidth. Preferably, the LO would be designed to only cover thefrequency regions of interest, that is the cellular bands. For thereference channel, the log amplifier will determine the peak or averagedpeak amplitude. The limiter output from the log amplifier will be passedthrough a comparator and into a counter that will determine the signalfrequency. On the measurement channel, an additional switchableattenuator will be used after the amplifier in order to increase therange of the allowable input signal strength. An RMS detector willmeasure the amplitude of the measurement channel. Optionally, the samedetector could also be used to determine the peak amplitude. Using bothdetectors, it is possible to detect and measure the signal strength ofthe received modulated RF energy with various modulation formats.

For the phase measurement, two phase detectors may be used. One willinput the reference and measurement channels directly from the IFfilters, while the other will have a 90 degree phase delay filter on thereference channel.

A microprocessor will control and read the measurements from theassociated A/D converters and the counter. It will communicate with theprocessing engine to determine the input frequency band and othernecessary information, and it will transmit the signal measurements tothe processing engine. To achieve the required precision, the A/Dconverters should preferably have a minimum of 10 bits of resolution.The sample rate should preferably be at least 1 MSPS, although having afaster sample rate will likely reduce the required time to make allrequired measurements and allow for some averaging of the data as well.

The amplitude and phase measured by the RF sampler is in a raw state towhich various corrections are applied to create an accurate data set ofthe scan plane. Initially, the RF sampler amplitude and phase correctionis applied at a given frequency and for a given temperature.Subsequently, path loss correction is applied to both amplitude andphase at a given frequency and for a given temperature. Lastly, thecorrected amplitude and phase data is converted to field quantities byapplying antenna factor correction.

As each element of the antenna array measures only one magnetic fieldcomponent orthogonal to that of its adjacent one, interpolation isapplied in order to get both transverse components at each samplingpoint of the scan plane. For amplitude, interpolation is implemented byaveraging its four adjacent measured points. For edge elements, data isinterpolated from its adjacent three elements. For corner elements, datais interpolated from its adjacent two elements. In one embodiment, phaseinterpolation may be achieved by a three points method. First, the 4adjacent data points are sorted from minimum to maximum. If the phasedifference between the sorted adjacent data points is larger than apre-determined threshold value, the most unique one is discarded and theremaining three points are averaged. Otherwise, the four adjacentmeasured points are averaged. Preferably, special treatments for edgepoints and 4 corners may be used in order to get better results.Alternatively, extrapolation from internal points is adopted for thosepoints.

In the amplitude and phase detection module (20), after the raw data ispassed through the correction and interpolation stages, the amplitudeand phase of the near field data is available for further processingdisplay and storage.

The mutual coupling effect, which may include effects due toreflections, dynamic coupling between individual antenna elements acrossthe array, and the DUT proximity effect, may be accounted for usingmethods described herein. As well, the finite scanner size is accountedfor using methods well known in the art. Computations are performed tocompute various models and their NF corrections. Far-field radiationpattern and radiated power of the antennas can be measured and studiedby measuring near-field radiation [Johnson J. H. Wang, “An Examinationof the Theory and Practices of Near-field Measurements,” IEEE Trans.Antennas Propagat., Vol. 36 pp. 746-753, January 1986].

FIG. 6 depicts the exterior fields of a radiating antenna, which arecommonly divided into three regions: reactive near-field region,radiating near-field region and far-field region. The reactivenear-field is excited in a small volume, just beyond the antenna andaccounts for the stored electric and magnetic energies around theantenna and attenuates very rapidly. The reactive near-field region iscommonly taken to extend about λ/2π from the surface of the antenna,although conventional near-field measurements use a distance of awavelength (λ) or greater to minimize the system uncertainty [Arthur D.Yaghjian, “An Overview of Near-field Antenna Measurements,” IEEE Trans.Antennas Propagat., Vol. AP-34 pp. 30-45, January 1986.]

Conventional scanning techniques of near-field measurement of antennasare based on the plane-wave spectrum (PWS) representation of fields thatcan be found in publications of Whittaker and Watson [G. T. Whittakerand G. N. Watson, Modern Analysis, 4^(th) ed. London: Cambridge Univ.Press, 1927, ch. XVIII].

A planar near-field measurement system is depicted in FIG. 7. Theaperture of the radiating antenna is in x-y plane of z≦0. The plane fornear-field measurement is in x-y plane of z=z_(t). Considering that theregion of z>0 is source-free, the solutions to the time-harmonicelectromagnetic field in front of the antenna aperture can be expressedas

$\begin{matrix}{{E\left( {x,y,z} \right)} = {\frac{1}{2\pi}{\int_{- \infty}^{+ \infty}{\int_{- \infty}^{+ \infty}{{A\left( {k_{x},k_{y}} \right)}^{{- j}\; {k \cdot r}}{k_{x}}{k_{y}}}}}}} & (1) \\{{H\left( {x,y,z} \right)} = {\frac{1}{2\pi}{\int_{- \infty}^{+ \infty}{\int_{- \infty}^{+ \infty}{k \times {A\left( {k_{x},k_{y}} \right)}^{{- j}\; {k \cdot r}}{k_{x}}{k_{y}}}}}}} & (2)\end{matrix}$with k _(x) A _(x)(k _(x) ,k _(y))+k _(y) A _(y)(k _(x) ,k _(y))+k _(z)A _(z)(k _(x) ,k _(y))=0  (3)

where k_(x) and k_(y) are real variables and k=k _(x) â _(x) +k _(y) â_(y) +k _(z) â _(z)  (4)

k may be called as wave number vector and A(k_(x),k_(y)) is called asthe plane wave spectrum because the expression A(k_(x), k_(y))e^(−jk·r)in the integrants represents a uniform plane wave propagating in thedirection k.

The equations are transformed and rearranged to express PWS A(k_(x),k_(y)) from Near-field using component H(x, y, z)

$\begin{matrix}{{{k_{y}{A_{z}\left( {k_{x},k_{y}} \right)}} - {k_{z}{A_{y}\left( {k_{x},k_{y}} \right)}}} = {^{j\; k_{z}z_{t}}\frac{1}{2\pi}{\int_{- \infty}^{+ \infty}{\int_{- \infty}^{+ \infty}{{H_{x}\left( {x,y,z_{t}} \right)}^{j{({{k_{x}x} + {k_{y}y}})}}{x}{y}}}}}} & (5) \\{{{k_{z}{A_{x}\left( {k_{x},k_{y}} \right)}} - {k_{x}{A_{z}\left( {k_{x},k_{y}} \right)}}} = {^{j\; k_{z}z_{t}}\frac{1}{2\pi}{\int_{- \infty}^{+ \infty}{\int_{- \infty}^{+ \infty}{{H_{y}\left( {x,y,z_{t}} \right)}^{j{({{k_{x}x} + {k_{y}y}})}}{x}{y}}}}}} & (6)\end{matrix}$

In the far-field zone of the antenna (kz>>1), based on the method ofsteepest descent, it can be shown that equation (1) can be representedby the asymptotic expansion [P. C. Clemmow, The Plane Wave SpectrumRepresentation of Electromagnetic Fields. London: Pergamon, 1966]

$\begin{matrix}{{E\left( {x,y,z} \right)} = {\frac{{j}^{{- j}\; {kr}}}{r}k_{z}{A\left( {k_{x},k_{y}} \right)}}} & (7)\end{matrix}$

When planar near field scanning is performed on a radiating surface, dueto the practical reasons and limitations, the scan has to be limited toa finite area in the x-y plane. Plane wave spectrum transformation maybe applied on this scanned data to determine the far field properties ofthe radiating surface. The accuracy of the far field transformed data ata given frequency is limited by the finite area used for scanning. Thedata may be further processed in a post-processing module to improve theaccuracy.

Conventional radiated power measurements are performed either in freespace or in presence of huge ground plane. The far field data estimatedusing PWS provides estimates in free space. The data set is corrected toaccount for the ground plane interactions, as necessary.

Calculations of Power Density Pattern or Radiation Pattern, DirectiveGain, Radiated Power and EIRP may be performed as follows:

${U\left( {\theta,\varphi} \right)} = {{{S\left( {\theta,\varphi} \right)}R^{2}} = {{\frac{1}{2}{{{Re}\left( {\overset{\rightarrow}{E} \times \overset{\rightarrow}{H}} \right)} \cdot \hat{r}}R^{2}} = {\frac{{E}^{2}R^{2}}{2Z} = {\frac{Z^{2}{H}^{2}R^{2}}{2Z} = {\frac{Z^{2}k^{2}}{2Z}\left\lbrack {{\left( {1 - a_{y}} \right)^{2}{M_{x}}^{2}} + {\left( {1 - a_{x}} \right)^{2}{M_{y}}^{2}} + {2a_{z}{a_{y}\left\lbrack \left( {{{{Re}\left( M_{x} \right)}{{Re}\left( M_{y} \right)}} + {{{Im}\left( M_{x} \right)}{{Im}\left( M_{y} \right)}}}\quad \right. \right\rbrack}}} \right\rbrack}}}}}$

The Z² is taken out as P_(offset), which will also takes othercoefficient into account. In Matlab, PDS→U. Radiated power is obtainedby integrating power density over the hemisphere. The hemisphere isdivided into 50×100 pieces. And again, integration is carried out bysumming the power density over the hemisphere.

For one complete scanning, the value can be obtained. If the scan iscontinued one after other, the quasi real time curve can be provided.

P _(rad)=∫₀ ^(2π)∫₀ ^(0.5π) U(θ,φ)sin θ·dθdφ

In the current implementation, dθ=1.8°, dφ=3.6°.

The power gain of an antenna in the direction specified by the sphericalco-ordinates (σ,φ) is defined as:

$\begin{matrix}{{{G_{p}\left( {\theta,\phi} \right)} = {4\pi \frac{U\left( {\theta,\phi} \right)}{P_{in}}}},} & (30)\end{matrix}$

where U(σ,φ), radiation intensity, is defined as “the power radiatedfrom an antenna per unit solid angle” [C. A. Balanis, “Antenna Theory:Analysis and Design”, Second Edition, John Wiley & Sons, 1997] in thedirection (σ,φ), and P_(a) is the total power accepted by the antennafrom the source. P_(in) is computed from the voltage and current at thesource as:

$\begin{matrix}{P_{in} = {\frac{1}{2}{{Re}\left( {V\; I^{*}} \right)}}} & (31) \\{and} & \; \\{{U\left( {\theta,\phi} \right)} = {{\frac{1}{2}R^{2}{{Re}\left( {E \times H^{*}} \right)}} = \frac{{E}^{2}R^{2}}{2\eta}}} & (32)\end{matrix}$

E is obtained from equation (28) with r in the direction (σ,φ), and r=R.Directivity is similarly defined as:

$D = {4\pi \frac{U\left( {\theta,\phi} \right)}{P_{rad}}}$

where P_(rad) is the total power radiated by the antenna,

$\begin{matrix}{P_{rad} = {P_{in} - P_{loss}}} \\{= {{∯_{\Omega}{U{\Omega}}} = {\int_{0}^{2\pi}{\int_{0}^{\pi}{U\; \sin \; \theta {\theta}{\phi}}}}}}\end{matrix}$

and P_(loss) is the total ohmic loss in the antenna.

If the direction is not specified, it implies the direction of maximumradiation intensity (maximum directivity) express as

$D_{\max} = {\frac{U_{\max}}{U_{0}} = {4\pi \frac{U_{\max}}{P_{rad}}}}$

The Effective Isotropic Radiated Power (EIRP) is the apparent powertransmitted towards the receiver, if it is assumed that the signal isradiated equally in all directions, i.e. as a spherical wave emanatingfrom a point source. This power is given by:

EIRP=G _(t) ·P _(t)

=D·P _(rad)

-   -   where:    -   Gt=gain of transmitter antenna,    -   Pt=power transmitted

EXAMPLES

The following examples are illustrative of the claimed invention, butnot limiting thereof.

The typical accuracies realizable in the industry for gain anddirectivity using far field measurement techniques is of the order of+/−0.25 dB over the cell phone operating frequency ranges. To achievethe traceability, extensive electromagnetic numerical simulation wereperformed to realize similar far field accuracies by realizing andadjusting the numerical model parameters of the reference sources at thepre defined cell phone band frequencies. Using these simulations, theEIRP of the reference sources was found to be 29.66 dBm and 24.95 dBmwith an accuracy of +/−0.3 dB at 1880 MHz and 836.4 MHz respectively.The near field amplitude and phase accuracies at very close distanceswere estimated from the near field data set derived from the far fieldsimulations and found to be of the order of 0.30 dB and +/−5 degrees.Using the amplitude and phase data from the simulations a frequency andmodel sensitive NF correction factor was developed to calibrate thescanner system to +/−0.3 dB amplitude and +/−5 degree phase accuracies.

FIG. 8A shows 3D near field total amplitude distribution of theradiating device under test. It is the resultant amplitude of the x andy magnetic filed intensities of the radiating device measured by eachprobe positioned at a predefined physical location.

FIG. 8B shows 2D near field amplitude distribution of x and y componentsof the radiating device under test. It is the amplitude of the x and ycomponents of magnetic filed intensities of the radiating devicemeasured by each probe positioned at a predefined physical location.

FIG. 8C shows the estimated value of EIRP, Directivity and RadiatedPower (real time display) of the radiating device. The Radiated power iscomputed from the corrected near field amplitude and phase distributionas well as applying appropriate near field to far field transformations.The directivity and EIRP are computed further from the radiated powerand computed radiation pattern of the radiating device.

FIG. 8D shows 3D Hemispherical Radiation Pattern of the radiating deviceand is computed after applying the near field to far fieldtransformations to corrected near field amplitude and phasedistributions.

FIG. 8E shows an integrated GUI combining FIGS. 8A, 8B, 8C and 8D. Anyon of these figures can be enlarged to show clearly the parameters thatare being displayed. The displays showed in FIGS. 8A and 8B can beinterchangeable by selecting appropriate options in the menu bar. Thetop right quadrant displays polar representation of the radiationpattern where the standard pattern of the device under test obtainedfrom any test laboratory can also be super imposed on the computedradiation pattern of the scanner system.

As will be apparent to those skilled in the art, various modifications,adaptations and variations of the foregoing specific disclosure can bemade without departing from the scope of the invention claimed herein.The various features and elements of the described invention may becombined in a manner different from the combinations described orclaimed herein, without departing from the scope of the invention.

1. An absorberless near field microwave scanning system comprising: (a)a switched array of antenna elements embedded in a dielectric, forsensing electromagnetic field components at pre-determined locations,and forming an array surface, wherein said array outputs raw uncorrectedsignals which are indicative of the electromagnetic field and whichinclude mutual coupling effects; (b) a scan surface for placement of thedevice under test (DUT), wherein the scan surface is substantiallyparallel to the array surface and separated by a distance less thanabout 1/1.8^(th) of the wavelength of the measured frequency; (c) aprocessing engine operatively connected to the switched antenna arrayfor obtaining and processing the array output, said processing engineadapted to correct at an individual probe level for mutual couplingeffects, and said processing engine comprising: i. a controller, ii. achannel selector and sampler, iii. a channel corrector, to adjust fordifferential path losses and delays iv. a data translator andinterpolator, v. an amplitude and phase detector, vi. a near fieldcorrector, vii. a transformer for transforming the near field data tofar field data, and ix. a user interface.
 2. The system of claim 1wherein the antenna elements are comprised in a multi-layer structure,and the antennas are isolated by means of ground vias through themulti-layer structure.
 3. The system of claim 1 wherein the distancebetween the scan surface and the array surface (D) is between about1/88^(th) to about 1/1.8^(th) of wavelength.
 4. The system of claim 1, 2or 3 wherein the inter element distance (d) range of the array isbetween about 1/176^(th) to about 1/3.6^(th) of the wavelength.
 5. Thesystem of claim 4 wherein D/d is about 2.0.
 6. The system of claim 1wherein the processing engine is further adapted to account for finitescanner size.
 7. A method of measuring and testing a performanceparameter of an electromagnetic radiating device, said method comprisingthe steps of: (a) using a switched array of antenna elements forming asubstantially parallel or non parallel array surface; (b) using thisarray scan surface, wherein the scan surface is substantially parallelto the array surface and separated by a distance less than about ½wavelength of the measured frequency; (c) generating near field data byreceiving output from the switched antenna array which is indicative ofan electromagnetic field but includes mutual coupling effects; (d)correcting the near field data to correct at an individual probe levelfor mutual coupling effects; and (e) transforming the corrected nearfield data to far field data.
 8. The method of claim 7 wherein theantenna elements are comprised in a multi-layer structure, and theantennas are isolated by means of ground vias through the multi-layerstructure.
 9. The method of claim 7 wherein the distance between thescan surface and the array surface (D) is between about 1/88^(th) toabout 1/1.8^(th) of wavelength.
 10. The method of claim 7 wherein theinter element distance (d) range of the array is between about1/176^(th) to about 1/3.6^(th) of the wavelength.
 11. The method ofclaim 9 or 10 wherein D/d is about 2.0.
 12. The method of claim 7wherein the mutual coupling effects corrected for include reflectionsand dynamic coupling effects between individual antenna elements acrossthe array, and DUT proximity to scanner surface.
 13. The method of claim12 wherein the near field data is further corrected to account forfinite scanner size.